The present invention relates to a negative feedback amplifier of a Cartesian loop system arranged to compensate (or correct) nonlinear distortion of a power amplifier for amplifying an orthogonally modulated signal of a transmitter and a transmitter itself, and more particularly to a method of compensating or correcting a phase error and an amplitude error of a quadrature demodulator to be used for the feedback loop.
The Cartesian loop negative feedback amplifier is an amplifier arranged to realize a negative feedback with signals orthogonal to each other. This amplifier is used as a digital radio communication having adopted a linear modulating system such as a π/4 shift QPSK modulating system or a hexadecimal QAM modulating system, specifically, as a power amplifier for compensating for nonlinear distortion of a transmitter in a narrow band digital ratio communication system. This sort of negative feedback amplifier is arranged to have a quadrature modulator and a quadrature demodulator, the electric performances of which modulator and demodulator, in particular, of the latter is likely to determine the overall performance of the transmitter.
Hence, when designing the negative feedback amplifier, high-precision circuit components have been conventionally used therefor. For example, refer to the thesis: “A Digital Cellular Equipment with Linear Modulation”, Shimazaki et. al, Transactions of Spring National Convention Record of IEICE (The Institute of Electronics, Information and Communication Engineers) in 1989, B-815. Later, the prior art will be described with reference to FIGS. 9 and 10.
FIG. 9 is a block diagram for illustrating the conventional negative feedback amplifier. At first, the description will expalin the operation of the negative feedback amplifier. In FIG. 9, an input baseband signal is generated by applying a predetermined digital modulating system to transmission data. Then, the in-phase component of the input baseband signal, that is, the I signal, and the quadrature component thereof, that is, the Q signal are passed through LPFs 4a and 4b, from which the transmission baseband signals Itx and Qtx are applied into adders 5a and 5b. Herein, a PN (Pseudo Noise) generator 90 is served as generating PN code series transmission data. A communication logic circuit 91 is served as converting the transmission data into the corresponding I and Q signals according to the predetermined communication format and modulating system and then outputting the I and Q signals to a D/A (digital-to-analog) converter (not shown).
On the other hand, the feedback baseband signals Id and Qd, which are outputted from a quadrature demodulator 13, are applied into adders 5a and 5b. The adders 5a and 5b operate to subtract the signals Id and Qd from the transmission baseband signals Itx and Qtx, respectively. That is, the adders 5a and 5b performs negative additions. The output signals of the adders 5a and 5b are applied into a quadrature modulator 7, in which the signals are quadrature-modulated with a local signal LO inputted from the other terminal, the local signal LO being outputted from a local oscillating circuit 9.
The quadrature modulator 7 is composed of a 90-degree phase shifter 31a, mixers 32a and 32b, and an adder 33. The 90-degree phase shifter 31a is inputted with the local signal LO (angular frequency: ω0 and supplies two local signals LOi (=cos ωot) and LOq (=sin ωot) whose phases are shifted by 90 degrees to each other. The mixers 32a and 32b operate to multiply the inputted Im and Qm signals by the local signals LOi and LOq, respectively and then upconvert them. Then, the upconverted signals are added in an adder 33, in which the added signal is made to be an orthgonally modulated signal, that is, a radio signal.
The radio signal, outputted from the quadrature modulator 7, is power-amplified in a power amplifier 8 and then is outputted from an output terminal 11. Ordinarily, the output terminal 11 is connected with an antenna (not shown), from which a radio wave is radiated.
A part of radio wave outputted from the power amplifier 8 is branched in a directional coupler 10. Then, the branched signal is inputted into the quadrature demodulator (also called a quadrature detector) 13. The quadrature demodulator 13 is composed of a 90-degree phase shifter 31b and two mixers 32c and 32d. The 90-degree phase shifter 31b is inputted with the local signal LO inputted from the other terminal, the local signal LO being outputted from the local oscillating circuit 9, and outputs two local signals LOi (=cos ωot) and LOq (=sin ωot) whose phases are shifted by 90 degrees to each other.
In the quadrature demodulator 13, the mixers 32c and 32d operate to multiply a part of radio wave by the local signals LOi and LOq, respectively. The multiplied signals are made to be the feedback baseband signals Id and Qd. Then, the feedback baseband signals Id and Qd are negatively fed back to the transmitting baseband signals Itx and Qtx in the adders 5a and 5b, respectively. This signal circulation completes the negative feedback loop by which a nonlinear distortion of the power amplifier 8 is compensated.
In turn, the description will explain the influence of phase and amplitude errors onto the transmission performance in the quadrature modulator and the quadrature demodulator. In FIG. 9, if the phases are not balanced with each other in the 90-degree phase shifter 31a and the 90-degree phase shifter 31b, there arises a phase error that the phase difference between the I-component and the Q-component is not just 90 degrees. Further, if the gains of the mixers 32a, 32b, 32c, 32d are not balanced with one another, there arises an amplitude error that the amplitude of the I-component is not matched to that of the Q-component. Herein, assuming that an ideal phase difference (90 degrees) between the two local signals LOi and LOq is as a reference, a phase error is represented by δ. Assuming that the I signal is a reference, an amplitude error of the Q signal is convergence in the I-Q signal space contained in the transmission wave to be shifted from an ideal point of convergence. This shift brings about a degrade of a sensitivity of a receiver having received this transmission wave.
For example, the description will be expanded with an example of a π/4 shift QPSK modulating system. In the I-Q signal space of FIG. 10, eight points indicated by circles represent ideal points of convergence on the I-Q space. (These eight points are ranged on the circumference of a unit circle at regular intervals of 45 degrees. In the quadrature modulator 7 or the quadrature demodulator 13, a phase unbalance (phase error: δ) is brought about in the 90-degree phase shifter 31a or 31b, thereby making LOq=sin(ωot+δ). This phenomenon appears on the I-Q space as follows. As shown in FIG. 10, a Q axis that is phase-shifted by 90 degrees from an I axis is rotated by δ so that the Q axis is made to be Qz. FIG. 10 shows an example of δ=10 degrees, at which the eight points of convergence are shifted from circles to triangles. The movement vector at this shift is called a residual vector error, and the effective value about the residual vector error of all points of convergence is called an error vector magnitude (EVM). The ideal points of convergence (circles) are ranged on a round circle, while the actual points of convergence (triangles) caused by the phase error δ are ranged on an inclined ellipse. This results in making the transmission performance (error vector magnitude) degraded. Further, the gains unbalanced among the mixers cause the amplitude error κ between the I signal and the Q signal to be added to the ellipse so that the ellipse is further distorted. This makes the signal more degraded.
In the conventional negative feedback amplifier, it is difficult to easily correct the phase error and the amplitude error. In order to overcome this shortcoming, the use of a highly accurate ring modulator having a wide band characteristic makes it possible to prevent the error vector magnitude from being degraded without correcting the errors.
The technologies about a Cartesian loop negative feedback amplifier have been disclosed in JP-A-2002-111759, JP-A-2001-339452, JP-A-10-136048, and JP-A-5-175743.
However, the foregoing prior arts involve the following shortcomings.
As a first shortcoming, in the case of using a microwave circuit like a ring modulator, it is less disadvantageous in light of reduction of the device in size and cost. In particular, if the radio frequency is low (for example, VHF band or lower), it is difficult to apply the device to a portable phone terminal. Hence, the prior art is required to use an IC (Integrated Circuit) of the commercially available quadrature modulator and demodulator though they are not so high in accuracy. It means that the error vector magnitude (EVM) of the transmitter depends on the electric performance of the commercially available IC. In order to improve the error vector magnitude, it is necessary to use an expensive highly accurate IC in place of the commercially available IC. Further, if such a highly accurate IC is not commercially available, it is necessary to newly develop a new IC dedicate therefor.
As a second shortcoming, even in the case of using the ring modulator or the IC for the modulator or the demodulator of the communication apparatus, it is not possible to compensate errors further degraded by some factors (aging, temperature change and so forth) after the product of the communication apparatus is delivered to the user from the factory. It means that the communication apparatus is required to perform periodic maintenance operations for correcting errors (for example, returning the product to the factory and compensating the errors at the factory).